Transmitting apparatus, receiving apparatus, transmitting method, receiving method, information recording medium and program

ABSTRACT

In a transmission device  101 , a modulation portion  102  carries out modulation of encoded data based on an adaptive modulation command based on feedback information sent from the receiving side, a frequency symbol diffusion block  105  multiplies the plurality of signals outputted by a serial-parallel conversion portion  104  by an orthogonal diffusion code and combines them, a pseudo random-number multiplication portion  106  multiplies each of them by a pseudo random number, an inverse Fourier transform portion  107  conducts inverse Fourier transform, a parallel-serial conversion portion  108  conducts parallel-serial conversion, a guard interval addition portion  109  adds a guard interval and a transmission portion  110  transmits a signal so that only one feedback information and modulation level information is required for each frequency symbol diffusion block  105  and transmission rate can be improved.

TECHNICAL FIELD

The present invention relates to a transmission device, receivingdevice, transmission method, receiving method, computer-readableinformation recording medium that records a program realizing them usinga computer, and the program suitable for improvement of performance ofadaptive OFDM (Adaptive Orthogonal Frequency Division Multiplexing)communication.

BACKGROUND ART

A demand for high data rate and high-quality multimedia service has beenraised in the radio communication field recently. In the mobile wirelessenvironment, signals are usually deteriorated by fading or multipathdelay phenomenon.

In such a communication channel, an influence of fading on amplitude ofa signal might become serious or an influence of inter-symbolinterference (ISI; Inter-Symbol Interference) might become serious byfrequency selectivity of the channel, which lowers error performance andmight disable communication depending on the case.

On the other hand, the OFDM technology is an effective method to reducethese influences of a multipath channel. That is because the ISI can beerased by inserting a guard interval longer than a delay spread of thechannel.

Thus, the OFDM is employed in various next-generation wide-area WLAN(Wireless Local Area Network) of IEEE 802.11a, IEEE 802.11g, EuropeanHIPERLAN/2 and the like.

Ground digital audio broadcasting (DAB; Digital Audio Broadcasting) anddigital video broadcasting are also proposed for the wide-area radiomultiple access system. They are IEEE 802.16 wireless MAN standard andinteractive DVB-T, for example.

Many of the OFDM systems use a fixed modulation scheme for all thecarriers; this is for simplification.

However, there is a possibility that performance is improved by using adifferent demodulation scheme according to a channel state for each subcarrier of the OFDM system.

In this case, coherent or differential phase- or amplitude modulationscheme may be used. It includes BPSK, QPSK, 8PSK, 16QAM, 64QAM and thelike, for example.

Each modulation scheme has a tradeoff between spectral efficiency andbit error rate (BER).

Thus, the best modulation scheme is such that the bit error rate is anallowable degree and the spectral efficiency can be maximized.

Such adaptive modulation schemes are disclosed in the documentsmentioned below:

Non-Patent Literature 1: C. Ahn and I. Sasase, The effects of modulationcombination, target BER, Doppler frequency, and adaptive interval on theperformance of adaptive OFDM in broadband mobile channel, IEEE Trans.Consumer Electronics, vol. 48, no. 1, pp. 167-174, February, 2002

Non-Patent Literature 2: T. Nakanishi, S. Sampei and N. Morinaga,Variable coding rate OFDM transmission on one-cell reuse TDMA systems,IEICE Trans. Communications, vol. EB-88, no. 2, pp. 535-540, February,2005

Non-Patent Literature 3: C. Ahn, S. Takahashi and H. Harada,Differential Modulated Pilot Symbol Assisted Adaptive OFDM for Reducingthe MLI with Predicted FBI, IEICE Trans. Communications, vol. EB-88, no.2, pp. 436-442, February, 2005

Non-Patent Literature 4: C. Ahn, S. Takahashi and H. Harada,Differential Modulated Pilot Symbol Assisted Adaptive OFDM for Reducintthe MLI, Proc. of IEEE TENCON 2004, pp. 577-580, Chiang Mai, Thailand,November, 2004

As disclosed in the [Non-Patent Literature 1], in the AdaptiveModulation Scheme (AMS)/OFDM system, it is necessary to control amodulation level for each sub carrier at base station according tofeedback information (FBI; Feedback Information).

The FBI includes evaluation results of channel state information (CSI)such as intensity and noise level of the respective sub carriers, forexample.

It is general, here, to assume that accuracy of the FBI is indefiniteand transmission of FBI can be ignored. However, in actual application,the transmission of FBI can be a serious problem.

Moreover, if an adaptive-modulated packet is to be transmitted from abase station to a mobile station after the base station controls themodulation level of each sub carrier, the mobile station needsmodulation level information (MLI) for demodulation of the receivedpacket.

Since the MLI is generally transmitted as data symbol, throughput ofdownlink of AMS/OFDM is deteriorated.

In the [Non-Patent Literature 2], such a scheme is proposed that a blockof the AMS/OFDM sub carrier is fixed and an encoding rate is madevariable for each block.

With this scheme, adjacent sub carriers are made into a block andassigned to the same modulation scheme among various encoding rates. Bythis arrangement, an amount of MLI transmission is reduced.

However, if the block size becomes large, the throughput is lowered bymismatch between the block modulation level and channel state.

Moreover, the number of required encoders and decoders is increased.

In the [Non-Patent Literature 3][Non-Patent Literature 4], apilot-symbol-assisted adaptive OFDM system in differential modulation(DMPSA-AMS/OFDM) is proposed so that the MLI transmission amount isreduced.

In the DMPSA-AMS/OFDM system, the MLI is transmitted as a pilot symboldifferentially modulated with FEC. Thus, the pilot symbol does not carryany information, and the transmission rate is not lowered.

However, delay time required for differentially demodulating anddecoding the received pilot symbol so as to obtain MLI becomes longer.

DISCLOSURE OF THE INVENTION Problem to be Solved by the Invention

It would be a practical system if the transmission amount of FBI and MLIcan be reduced with respect to the AMS/OFDM.

This application has an object to provide a transmission device,receiving device, transmission method, receiving method, acomputer-readable information recording medium recording a program thatrealizes them using a computer, and the program that solves the aboveproblem and improves total throughput of adaptive OFDM communication.

Means for Solving the Problem

In order to achieve the above object, the following invention will bedisclosed according to the principle of the present invention.

A transmission device according to a first aspect of the presentinvention comprises a serial-parallel conversion portion, frequencysymbol diffusion portion, pseudo random-number multiplication portion,inverse Fourier transform portion, parallel-serial conversion portion,and transmission portion and configured as follows.

Here, the serial-parallel conversion portion serial-parallel converts atransmission signal to Nc pieces and outputs a plurality of signals, andthe i-th symbol in the time direction of the n-th signal in theplurality of signals is:d(n,i).

On the other hand, the frequency symbol diffusion portion outputs aplurality of signals using a complex orthogonal diffusion seriesc_(k)(m) with the length of Nsf with respect to the outputted pluralityof signals d(n,i). Here,|c _(k)(m)|=1and if k=w, it isΣ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=Nsf;if k≠w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=0and (·)* acquires complex conjugation and floor (·) conducts truncation.Among the plurality of signals, the i-th symbol in the time direction ofthe n-th signal is:u(n,i)=Σ_(k=0) ^(Nsf−1) c _(k)(n mod Nsf)·d(floor(n/Nsf)·Nsf+k,i).

Moreover, the pseudo random-number multiplication portion multiplieseach of the output plurality of signals u(n,i) byc_(PN)(n)out of the pseudo random-number code seriesc_(PN)(0),c_(PN)(1), . . .and outputs the result.

Then, the inverse Fourier transform portion conducts inverse Fouriertransform of the outputted plurality of signals c_(PN)(n)·u(n,i) andoutputs a plurality of signals.

On the other hand, the parallel-serial conversion portionparallel-serial converts the plurality of signals outputted after theinverse Fourier transform.

Moreover, the transmission portion transmits the signal of the result ofparallel-serial conversion.

A receiving device according to another aspect of the present inventioncommunicates with the transmission device and is provided with areceiving portion, serial-parallel conversion portion, Fourier transformportion, pseudo random-number multiplication portion, weight calculationportion, detection portion, frequency equalization combination portion,and parallel-serial conversion portion, which are configured as follows.

Here, the receiving portion receives a signal transmitted from thetransmission device.

On the other hand, the serial-parallel conversion portionserial-parallel converts the received signal to Nc pieces and outputs aplurality of signals.

Moreover, the Fourier transform portion conducts Fourier transform ofthe plurality of serial-parallel converted and outputted signals andoutputs a plurality of signals, and the i-th symbol in the timedirection of the n-th signal among the plurality of Fourier-transformedand outputted signals is:r{tilde over ( )}(n,i).

Then, the pseudo random-number multiplication portion multiplies each ofthe plurality of Fourier-transformed and outputted signals r{tilde over( )}(n,i) by complex conjugationc_(PN)(n)*ofc_(PN)(n)among the pseudo random-number code series and outputs it.

On the other hand, the weight calculation portion calculates a weight tothe i-th symbol of the n-th signal:w(n,i).

Moreover, the detection portion multiplies the plurality of signalsmultiplied by the complex conjugation c_(PN)(n)* and outputted by thecalculated weight w(n,i) and outputs a plurality of signals:u^(n,i)=w(n,i)·c _(PN)(n)*·r{tilde over ( )}(n,i).

Then, the frequency equalization and combination portion performsfrequency equalization and combination to the outputted plurality ofsignals u(n,i) and outputs a plurality of signals, and the i-th symbolin the time direction of the n-th signal in the plurality of signals is:d{tilde over ( )}(n,i)=Σ_(k=0) ^(Nsf−1) u(floor(n/Nsf)·Nsf+k,i)·c_(n mod Nsf)(k)*.

On the other hand, the parallel-serial conversion portionparallel-serial converts the outputted a plurality of signals d{tildeover ( )}(n,i) and obtains a transmission signal.

Also, the receiving device of the present invention is further providedwith a channel transfer function calculation portion, which may beconfigured as follows.

That is, the channel transfer function calculation portion calculates,using a pilot signal p(n,i) with intensity P, length Np transmitted fromthe transmission device, a channel transfer function H{tilde over ()}(n/Ts) by:H{tilde over ( )}(n/Ts)=1/(Np·(2P/Nc)^(1/2))Σ_(i=0) ^(Np−1) r{tilde over( )}(n,i)·p(n,i)*·c _(PN) i*

On the other hand, the weight w(n,i) is determined from the channeltransfer function H{tilde over ( )}(n/Ts).

Also, at the receiving device of the present invention, it may be soconfigured that the weight w(n,i) is determined as:w(n,i)=1/H{tilde over ( )}(n/Ts).

Also, at the receiving device of the present invention, by an averageσ{tilde over ( )}² of noise intensity evaluated for each of theplurality of signals r{tilde over ( )}(n,i), it may be so configuredthat the weight w(n,i) is determined as:w(n,i)=(2S/Nc)^(1/2) ·H{tilde over ( )}(n/Ts)/(|(2S/Nc)^(1/2) ·H{tildeover ( )}(n/Ts)|²+2σ²).

A transmission method according to another aspect of the presentinvention is provided with a serial-parallel conversion process,frequency symbol diffusion process, pseudo random-number multiplicationprocess, inverse Fourier transform process, parallel-serial conversionprocess, and transmission process, which are configured as follows.

Here, in the serial-parallel conversion process, a transmission signalis serial-parallel converted to Nc pieces and a plurality of signals areoutputted, and the i-th symbol in the time direction of the n-th signalin the plurality of signals is:d(n,i).

On the other hand, in the frequency symbol diffusion process, aplurality of signals are outputted using a complex orthogonal diffusionseries c_(k)(m) with the length of Nsf with respect to the outputtedplurality of signals d(n,i). Here,|c _(k)(m)|=1and if k=w, it isΣ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=Nsf;if k≠w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=0and (·)* acquires complex conjugation and floor (·) conducts truncation.Among the plurality of signals, the i-th symbol in the time direction ofthe n-th signal is:u(n,i)=Σ_(k=0) ^(Nsf−1) c _(k)(n mod Nsf)·d(floor(n/Nsf)·Nsf+k,i).

Moreover, in the pseudo random-number multiplication process, each ofthe outputted plurality of signals u(n,i) is multiplied byc_(PN)(n)out of the pseudo random-number code seriesc_(PN)(0),c_(PN)(1), . . .and outputs the result.

Then, in the inverse Fourier transform process, inverse Fouriertransform is conducted for the outputted plurality of signalsc_(PN)(n)·u(n,i) and a plurality of signals are outputted.

On the other hand, in the parallel-serial conversion process, theinverse-Fourier-transformed and outputted plurality of signals areparallel-serial converted.

Moreover, in the transmission process, the signal of the result of theparallel-serial conversion is transmitted.

A receiving method according to another aspect of the present inventionreceives a signal by the transmission method and is provided with areceiving process, serial-parallel conversion process, Fourier transformprocess, pseudo random-number multiplication process, weight calculationprocess, detection process, frequency equalization and combinationprocess, and parallel-serial conversion process, which are configured asfollows.

Here, in the receiving process, the signal transmitted by thetransmission method is received.

On the other hand, in the serial-parallel conversion process, thereceived signal is serial-parallel converted to Nc pieces and aplurality of signals are outputted.

Moreover, in the Fourier transform process, the plurality ofserial-parallel converted and outputted signals are Fourier-transformedand a plurality of signals are outputted, and the i-th symbol in thetime direction of the n-th signal among the plurality ofFourier-transformed and outputted signals is:r{tilde over ( )}(n,i).

Then, in the pseudo random-number multiplication process, each of theplurality of Fourier-transformed and outputted signals r{tilde over ()}(n,i) is multiplied by complex conjugationc_(PN)(n)*ofc_(PN)(n)among the pseudo random-number code series and outputted.

On the other hand, in the weight calculation process, a weight to thei-th symbol of the n-th signal:w(n,i)is calculated.

Moreover, in the detection process, the plurality of signals multipliedby the complex conjugation c_(PN)(n)* and outputted is multiplied by thecalculated weight w(n,i) and a plurality of signals:u^(n,i)=w(n,i)·c _(PN)(n)*·r{tilde over ( )}(n,i)are outputted.

Then, in the frequency equalization and combination process, frequencyequalization and combination is performed to the outputted plurality ofsignals u(n,i) and a plurality of signals are outputted, and the i-thsymbol in the time direction of the n-th signal in the plurality ofsignals is:d{tilde over ( )}(n,i)=Σ_(k=0) ^(Nsf−1) u(floor(n/Nsf)·Nsf+k,i)·c_(n mod Nsf)(k)*.

On the other hand, in the parallel-serial conversion process, theoutputted plural signals d{tilde over ( )}(n,i) are parallel-serialconverted so as to obtain a transmission signal.

Also, the receiving method of the present invention is further providedwith a channel transfer function calculation process, which may beconfigured as follows.

That is, in the channel transfer function calculation process, using apilot signal p(n,i) with intensity P, length Np transmitted from thetransmission device, a channel transfer function H{tilde over ( )}(n/Ts)is calculated by:H{tilde over ( )}(n/Ts)=1/(Np·(2P/Nc)^(1/2))Σ_(i=0) ^(Np−1) r{tilde over( )}(n,i)·p(n,i)*·c _(PN) i*

On the other hand, the weight w(n,i) is determined from the channeltransfer function H{tilde over ( )}(n/Ts).

Also, in the receiving method of the present invention, it may be soconfigured that the weight w(n,i) is determined as:w(n,i)=1/H{tilde over ( )}(n/Ts).

Also, in the receiving method of the present invention, by an averageσ{tilde over ( )}² of noise intensity evaluated for each of theplurality of signals r{tilde over ( )}(n,i), it may be so configuredthat the weight w(n,i) is determined as:w(n,i)=(2S/Nc)^(1/2) ·H{tilde over ( )}(n/Ts)/(|(2S/Nc)^(1/2) ·H{tildeover ( )}(n/Ts)|²+2σ²).

A program according to another aspect of the present invention ischaracterized in that a computer is configured to function as eachportion of the transmission device or each portion of the receivingdevice.

A computer-readable information recording medium according to anotheraspect of the present invention is configured to record the aboveprogram. The program may be recorded in a computer-readable informationstorage medium such as compact disk, flexible disk, hard disk,magneto-optical disk, digital video disk, magnetic tape, semiconductormemory and the like.

If the communicating device is configured using a computer or softwareradio technology using DSP (Digital Signal Processor) and FPGA (FieldProgrammable Gate Array), for example, the transmission device andreceiving device of the present invention is realized by executing theabove program, and the program may be distributed/sold to thecommunicating device via a computer communication network. Also, theinformation storage medium may be distributed/sold independently of thecommunicating device.

Effect of the Invention

According to the present invention, the transmission device, receivingdevice, transmission method, receiving method, computer-readableinformation recording medium recording the program realizing them usinga computer suitable for improvement of performance of adaptive OFDMcommunication, and the program can be provided.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is an explanatory diagram illustrating a schematic configurationof a transmission device of this embodiment.

FIG. 2 is an explanatory diagram illustrating a schematic configurationof a frequency symbol diffusion block.

FIG. 3 is an explanatory diagram illustrating intensity spectrums of aninput signal and an output signal of the frequency symbol diffusionblock.

FIG. 4 is an explanatory diagram illustrating a schematic configurationof a receiving device of this embodiment.

FIG. 5 is an explanatory diagram illustrating a state of intensity of asub carrier.

FIG. 6 is an explanatory diagram illustrating a state of transferchannel propagation received by a transmission signal.

FIG. 7 is an explanatory diagram illustrating a packet structure.

FIG. 8 is a graph illustrating a BER value to the conventional OFDM anda BER value to FSS-OFDM using ORC and MMSEC.

FIG. 9 is a graph illustrating a BER value to the conventional OFDM anda BER value to FSS-OFDM using ORC and MMSEC.

FIG. 10 is a graph illustrating a BER value to the conventional OFDM anda BER value to FSS-OFDM using ORC and MMSEC.

FIG. 11 is a graph illustrating throughputs of fixed QPSK OFDM, fixed16QAM OFDM, conventional AMS/OFDM, AMS/FSC-OFDM with ORC, AMS/FSC-OFDMwith MMSEC.

FIG. 12 is a graph illustrating throughputs of fixed QPSK OFDM, fixed16QAM OFDM, conventional AMS/OFDM, AMS/FSC-OFDM with ORC, AMS/FSC-OFDMwith MMSEC.

EXPLANATION OF REFERENCE NUMERALS

-   -   101 TRANSMISSION DEVICE    -   102 MODULATION PORTION    -   103 MULTIPLEXER    -   104 SERIAL-PARALLEL CONVERSION PORTION    -   105 FREQUENCY SYMBOL DIFFUSION BLOCK    -   106 PSEUDO RANDOM-NUMBER MULTIPLICATION PORTION    -   107 INVERSE FOURIER TRANSFORM PORTION    -   108 PARALLEL-SERIAL CONVERSION PORTION    -   109 GUARD INTERVAL ADDITION PORTION    -   110 TRANSMISSION PORTION    -   401 RECEIVING DEVICE    -   402 RECEIVING PORTION    -   403 GUARD INTERVAL REMOVAL PORTION    -   404 SERIAL-PARALLEL CONVERSION PORTION    -   405 FOURIER TRANSFORM PORTION    -   406 PSEUDO RANDOM-NUMBER MULTIPLICATION PORTION    -   407 DETECTION PORTION    -   408 CHANNEL EVALUATION PORTION    -   409 PARALLEL-SERIAL CONVERSION PORTION    -   410 DECODER

BEST MODE FOR CARRYING OUT THE INVENTION

An embodiment of the present invention will be described below. Theembodiment described below is for explanation and does not limit thescope of the present invention. Therefore, any embodiment in which eachor all the elements are replaced by equivalent elements by those skilledin the art may be employed, and these embodiments are also included inthe scope of the present invention.

Example 1

In configuration described below, the frequency symbol diffusion andMMSEC equalization are carried out based on an adaptive downlink OFDMsystem.

Here, on the transmission side, each of Nsf=N_(SF) pieces ofserial-parallel converted signals is diffused by an orthogonal diffusioncode with the length of Nsf and then, combined.

By this arrangement, on each sub carrier, a plurality of signalsserial-parallel converted with the same intensity rate are superimposed.

In this case, the sub carrier subject to an influence of frequencyselective fading is obtained with the same intensity rate for each ofthe plurality of serial-parallel converted signals.

Therefore, the same modulation level can be assigned to each frequencysymbol diffusion block. As a result, a detected signal can be obtainedalso with the same SINR.

Moreover, since SINR of each sub carrier presents the same value in thesame frequency symbol diffusion block, there is only one piece of FBIand MLI unless they are transmitted for each block. This is opposite theconventional AMS/OFDM.

As mentioned above, in the OFDM system described below, a transmissionamount of FBI and MLI can be reduced.

However, orthogonality between different diffusion codes might be lostby the frequency selective fading.

Then, in the present application, in order to restore the orthogonality,various frequency equalization technologies are proposed. For example,they include Orthogonal Restoration Combining (ORC) and Minimum MeanSquare Error Combining (MMSEC).

Details will be described below.

(Channel Model)

Suppose that a propagation channel consists of L pieces of discretepaths and the respective time delays are different below. Then, animpulse response h(τ,t) can be represented as in [Formula 1]:

$\begin{matrix}{{{h\left( {\tau,t} \right)} = {\sum\limits_{l = 0}^{L - 1}{{h_{l}(t)}{\delta\left( {\tau - \tau_{l}} \right)}}}},} & \left\lbrack {{Formula}\mspace{14mu} 1} \right\rbrack\end{matrix}$

Here, h_(l) and τ₁ are a complex channel gain and time delay of thefirst propagation path, respectively. If E|·| is a calculation toacquire an average, the following is true:Σ_(l=0) ^(L−1) E|h _(l) ²|=1

The channel transfer function H(f,t) is Fourier transform of h(τ,t) andcan be obtained as in [Formula 2].

$\begin{matrix}\begin{matrix}{{H\left( {f,t} \right)} = {\int_{0}^{\infty}{{h\left( {\tau,t} \right)}{\exp\left( {{- {j2\pi}}\; f\;\tau} \right)}{\mathbb{d}\tau}}}} \\{= {\sum\limits_{l = 0}^{L - 1}{{h_{l}(t)}{{\exp\left( {{- j}\; 2\;\pi\; f\;\tau_{l}} \right)}.}}}}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 2} \right\rbrack\end{matrix}$

In radio transmission, a channel spectrum response is not flat. In thecase of L>1, H(f,t) is not a constant on a signal band width.

Such a channel is called frequency selective fading channel, and thiswill be considered below with the purpose of evaluating adaptivedownlink FSS-OFDM system.

(Transmission Device)

FIG. 1 is an explanatory diagram illustrating schematic configuration ofa transmission device according to the adaptive downlink FSS-OFDM/TDMAsystem of this embodiment. This will be described below referring to thefigure.

A transmission device 101 comprises an encoder 111, a modulation portion102, a multiplexer 103, a serial-parallel conversion portion 104, afrequency symbol diffusion block 105, a pseudo random-numbermultiplication portion 106, an inverse Fourier transform portion 107, aparallel-serial conversion portion 108, a guard interval additionportion 109, and a transmission portion 110.

Here, a transmission signal is encoded by the encoder 111 and modulatedby the modulation portion 102 by a modulation method specified by anadaptive modulation command (AMC; Adaptive Modulation Command) generatedbased on the feedback information sent from the receiving device. Themultiplexer 103 adds Np pieces of pilot symbols to the beginning of themodulated signal string and multiplexes them.

The serial-parallel conversion portion 104 serial-parallel coverts thissignal and outputs Nc pieces of parallel signals.

The outputted Nc pieces of parallel signals are grouped (blocked) forNsf=N_(SF) pieces, and each block is given to the frequency symboldiffusion block 105.

Specifically, the n-th parallel signal is given to the (n−1) modN_(SF)-th sub code processing block of the floor(n/N_(SF))-th frequencysymbol diffusion block 105.

Here, Nsf=N_(SF) is a diffusion code length, floor(·) is truncationcalculation, and x mod y is calculation to obtain a residue when x isdivided by y.

floor(·) can be expressed by noting an expression requiring truncationbetween the one in which a side is drawn from up to down and a side isfurther drawn to the right at a right angle and the one in which a sideis drawn from up to down and a side is further drawn to the left at aright angle (Gaussian symbol). That is, floor(x) returns the maximuminteger not exceeding x.

FIG. 2 is an explanatory diagram illustrating schematic configuration ofthe frequency symbol diffusion block. This will be described belowreferring to the figure.

When a block of parallel signals is given to the frequency symboldiffusion block 105, the parallel signals are copied in the same numberas the length of an orthogonal diffusion code with the length of N_(SF),respectively.

The copied complex string is diffused by N_(SF) pieces of orthogonaldiffusion codes, respectively, and combined.

This state will be described below in more detail. As shown in thefigure, suppose that i₀, . . . , i_(k), . . . , i_(Nsf−1) are given asinput to each frequency symbol diffusion block 105 and outputs are o₀ .. . , o_(k), . . . , o_(Nsf−1).i₀ is copied and each is multiplied by c₀(0), . . . ,c₀(k), . . .,c₀(Nsf−1) respectively.. . .i_(k) is copied and each is multiplied by c_(k)(0), . . . ,c_(k)(k), . .. ,c_(k)(Nsf−1) respectively.. . .i_(Nsf−1) is copied and each is multiplied by c_(Nsf−1)(0), . . .,c_(Nsf−1)(k), . . . ,c_(Nsf−1)(Nsf−1) respectively.i₀c_(o)(0)+ . . . +i_(k)c_(k)(0)+ . . . +i_(Nsf−1)c_(Nsf1)(0) becomesoutput o₀.. . .i₀c_(o)(k)+ . . . +i_(k)c_(k)(k)+ . . . +i_(Nsf−1)c_(Nsf1)(k) becomesoutput o_(k).. . .i₀c_(o)(Nsf−1)+ . . . +i_(k)c_(k)(Nsf−1)+ . . .+i_(Nsf−1)c_(Nsf1)(Nsf−1) becomes output o_(Nsf−1).

Returning to FIG. 1, this relation will be further examined. If the i-thsymbol in the time direction of the n-th parallel signal isd(n,i)and |d(n,i)|=1, the combined result of the signal u(n,i) can beexpressed as in [Formula 3].

$\begin{matrix}{{{u\left( {n,{\mathbb{i}}} \right)} = {\sum\limits_{k = 0}^{N_{SF} - 1}{{c_{k}\left( {n\;{mod}\; N_{SF}} \right)} \cdot {d\left( {{{\left\lfloor {n/N_{SF}} \right\rfloor \cdot N_{SF}} + k},{\mathbb{i}}} \right)}}}},} & \left\lbrack {{Formula}\mspace{14mu} 3} \right\rbrack\end{matrix}$

This can be expressed as:u(n,i)=Σ_(k=0) ^(Nsf−1) c _(k)(n mod Nsf)·d(floor(n/Nsf)·Nsf+k,i).

Here, c_(k)(m) is a orthogonal diffusion series, which satisfies:|c _(k)(m)|=1and also satisfies [Formula 4].

$\begin{matrix}{{\sum\limits_{m = 0}^{N_{SF} - 1}{{c_{k}(m)}{c_{w}^{*}(m)}}} = \left\{ \begin{matrix}N_{SF} & {{{for}\mspace{14mu} k} = w} \\0 & {{{for}\mspace{14mu} k} \neq w}\end{matrix} \right.} & \left\lbrack {{Formula}\mspace{14mu} 4} \right\rbrack\end{matrix}$

If k=w, this can be written as;Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=Nsf;if k≠w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·*c _(w)(m)*=0

where ·* is calculation to acquire complex conjugation.

To the combined parallel signal obtained as above, the pseudorandom-number multiplication portion 106 is diffused in a frequencydomain by a long pseudo random-number scramble code:c_(PN)(0),c_(PN)(1), . . .

That is, by multiplying the signal u(n,i) by c_(PN)(n), diffusion isconducted.

After that, the inverse Fourier transform portion 107 carries outinverse fast Fourier transform (IFFT; Inverse Fast Fourier Transform).By this arrangement, the FSS-OFDM/TDMA signal waveform to be transmittedis obtained.

Moreover, the parallel-serial conversion portion 108 conductsparallel-serial conversion, the guard interval addition portion 109 addsguard interval and transmits a signal by the transmission portion 110made of an antenna.

The downlink FSS-OFDM/TDMA transmission signal can be expressed as[Formula 5] in an equivalent baseband expression:

$\begin{matrix}{{{s(t)} = {\sum\limits_{i = 0}^{N_{p} + N_{d} - 1}{{g\left( {t - {{\mathbb{i}}\; T}} \right)} \cdot \begin{Bmatrix}{\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{n = 0}^{N_{c} - 1}{{{c_{PN}(n)} \cdot u}{\left( {n,{\mathbb{i}}} \right) \cdot}}}} \\{\exp\left\lbrack {{{j2\pi}\left( {t - {{\mathbb{i}}\; T}} \right)}{n/T_{s}}} \right\rbrack}\end{Bmatrix}}}},} & \left\lbrack {{Formula}\mspace{14mu} 5} \right\rbrack\end{matrix}$

Here, Ts is an effective symbol length, S is an average transmissionintensity, and T is an OFDM symbol length. An interval of the adjacentorthogonal sub carrier frequencies is 1/Ts.

A guard interval with a length of Tg is inserted in order to eraseinter-carrier interference caused by the frequency selective fading.Therefore, [Formula 6] is true:T=T _(s) +T _(g)  [Formula 6]

From [Formula 5], [Formula 6], a transmission pulse is obtained as in[Formula 7]:

$\begin{matrix}{{g(t)} = \left\{ \begin{matrix}1 & {{- T_{g}} \leq t \leq T_{s}} \\0 & {otherwise}\end{matrix} \right.} & \left\lbrack {{Formula}\mspace{14mu} 7} \right\rbrack\end{matrix}$

FIG. 3 are explanatory diagrams illustrating intensity spectrums of aninput signal and an output signal of the frequency symbol diffusionblock, which will be described below referring to the figures.

FIG. 3A is a power spectrum of the input signal and FIG. 3B is a powerspectrum of the output signal.

As mentioned above, the parallel signal d(n,i) is given to thefloor(n/Nsf)-th frequency symbol diffusion block 105.

The input data d(n,i) is copied at a magnification of the Nsf times atone frequency symbol diffusion block 105 and multiplied (n mod Nsf)times. At the same frequency symbol diffusion block 105, the outputdiffusion signals are combined. Therefore, all the data is combined inthe frequency domain.

As shown in FIG. 3B, energy of the input data is divided by a diffusionsub code to Nsf pieces of sub carriers, and each sub carrier includesNsf pieces of divided data.

In this case, the diffusion data is given frequency diversity withoutchanging (increasing) intensity of each sub carrier.

(Receiving Device)

Outline of operation at the receiving device is as follows. That is,when an OFDM waveform is received, it is separated to Nc pieces oforthogonal sub carriers by applying fast Fourier transform (FFT), andthe transmitted data is obtained by inverse diffusion of the orthogonalsub carrier received by the orthogonal diffusion code and a scramblecode.

In the frequency selective fading, in the case of corruption of theorthogonality between diffusion codes with a possibility of corruption,in order to compensate it, the frequency equalization method such as ORCand MMSEC is used below at detection.

Detailed description will be given below. FIG. 4 is an explanatorydiagram illustrating schematic configuration of the receiving deviceaccording to this embodiment. Description will be made below referringto this figure.

A receiving device 401 is provided with a receiving portion 402, a guardinterval removal portion 403, a serial-parallel conversion portion 404,a Fourier transform portion 405, a pseudo random-number multiplicationportion 406, a detection portion 407, a channel evaluation portion 408,a parallel-serial conversion portion 409, and a decoder 410.

For a signal r(t) received through the receiving portion 402 made of anantenna, the guard interval removal portion 403 removes guard interval,the serial-parallel conversion portion 404 conducts serial-parallelconversion, and the Fourier transform portion 405 applies fast Fouriertransform to it so as to disassemble it to Nc pieces of sub carriers.

The receiving signal is frequency-equalized in order to reduce frequencydistortion caused by the frequency selective fading. Since thetransmission data symbol is obtained by multiplication of the orthogonaldiffusion code on Nc pieces of the sub carriers, the receiving signalr(t) can be expressed as [Formula 8] in equivalent baseband expression:r(t)=∫_(−∞) ^(∞) h(τ,t)s(t−τ)dτ+n(t)  [Formula 8]

Here, n(t) is an additive white Gaussian noise (AWGN: Additive WhiteGaussian Noise) of a one-side power spectral density N₀.

Then, the n-th sub carrier r{tilde over ( )}(n,i) is given as in[Formula 9]:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{r}\left( {n,{\mathbb{i}}} \right)} = {\frac{1}{T_{s}}{\int_{iT}^{{iT} + T_{s}}{{r(t)}{\exp\left\lbrack {{- {{j2\pi}\left( {t - {{\mathbb{i}}\; T}} \right)}}{n/T_{s}}} \right\rbrack}{\mathbb{d}t}}}}} \\{= {\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{e = 0}^{N_{c} - 1}{{{u\left( {e,{\mathbb{i}}} \right)} \cdot \frac{1}{T_{s}}}{\int_{0}^{T_{s}}{{\exp\left\lbrack {{{j2\pi}\left( {{\mathbb{e}} - n} \right)} \cdot {t/T_{s}}} \right\rbrack} \cdot}}}}}} \\{{\left\{ {\int_{- \infty}^{\infty}{{h\left( {\tau,{t + {{\mathbb{i}}\; T}}} \right)}{{g\left( {t - \tau} \right)} \cdot {\exp\left( {{- 2}\pi\;{\mathbb{e}}\;{\tau/T_{s}}} \right)}}{\mathbb{d}\tau}}} \right\}{\mathbb{d}t}} + {\hat{n}\left( {n,{\mathbb{i}}} \right)}}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 9} \right\rbrack\end{matrix}$

Here, n(n,i) is AWGN with an average 0, variance 2N₀/Ts.

Here, if the maximum τ_(l) is shorter than the guard interval length Tg,integration of τ is obtained as in [Formula 10] from [Formula 7].

$\begin{matrix}\begin{matrix}{{\int_{- \infty}^{\infty}{{h\left( {\tau,{t + {{\mathbb{i}}\; T}}} \right)}{g\left( {t - \tau} \right)}{\exp\left( {{- {j2\pi}}\;{\mathbb{e}}\;{\tau/T_{s}}} \right)}{\mathbb{d}\tau}}} = {\int_{0}^{T_{s}}{h\left( {\tau,{t + {{\mathbb{i}}\; T}}} \right)}}} \\{{\exp\left( {{- {j2\pi}}\;{\mathbb{e}}\;{\tau/T_{s}}} \right)}{\mathbb{d}\tau}} \\{= {{H\left( {{{\mathbb{e}}/T_{s}},{t + {{\mathbb{i}}\; T}}} \right)}.}}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 10} \right\rbrack\end{matrix}$

Here, suppose that ε_(i)(t) is approximately a constant on the symbollength T. That is, suppose as [Formula 11]:ε_(i)(t+iT)≈ε_(i)(iT) for 0≦t≦T  [Formula 11]

Then, [Formula 12] is obtained:H(n/T _(s) ,t+iT)≈H(n/T _(s) ,iT) for 0≦t≦T  [Formula 12]

As a result, [Formula 9] can be written as [Formula 13]:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{r}\left( {n,{\mathbb{i}}} \right)} \approx {\frac{1}{T_{s}}\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{e = 0}^{N_{c} - 1}{{u\left( {{\mathbb{e}},{\mathbb{i}}} \right)} \cdot}}}} \\{{\int_{0}^{T_{s}}{\exp\left\lbrack {{{j2\pi}\left( {{\mathbb{e}} - n} \right)} \cdot {t/T_{s}}} \right\rbrack}} + {\hat{n}\left( {n,{\mathbb{i}}} \right)}} \\{= {{\sqrt{\frac{2S}{N_{c}}}{H\left( {{n/T_{s}},{{\mathbb{i}}\; T}} \right)}{u\left( {n,{\mathbb{i}}} \right)}} + {\hat{n}\left( {n,{\mathbb{i}}} \right)}}}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 13} \right\rbrack\end{matrix}$

Referring to [Formula 13], it is known that there is frequencydistortion in the receiving signal caused by the frequency selectivefading. In order to reduce the frequency distortion, frequencyequalization and combination is required. Therefore, a weight, whichwill be described later, is used.

After the fast Fourier transform, c_(PN)(n)* is multiplied by the pseudorandom-number multiplication portion 406 for the n-th sub carrierr{tilde over ( )}(n,i).

Moreover, at the detection portion 407, the frequency equalization andcombination shown in [Formula 14] is carried out using the weightw(n,i).

$\begin{matrix}{{\overset{\sim}{d}\left( {n,{\mathbb{i}}} \right)} = {\sum\limits_{k = 0}^{N_{SF} - 1}{{\hat{u}\left( {{{\left\lfloor {n/N_{SF}} \right\rfloor \cdot N_{SF}} + k},{\mathbb{i}}} \right)}{c_{n\mspace{20mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}}}} & \left\lbrack {{Formula}\mspace{14mu} 14} \right\rbrack\end{matrix}$

Here, for k=0, 1, . . . , Nsf−1,u^(q+k,i)is a weighted element of the n-th sub carrier and can be expressed as in[Formula 15]:

$\begin{matrix}\begin{matrix}{{\hat{u}\left( {n,{\mathbb{i}}} \right)} = {{w\left( {n,{\mathbb{i}}} \right)}{c_{PN}^{*}(n)}{\overset{\sim}{r}\left( {n,{\mathbb{i}}} \right)}}} \\{= {{\sqrt{\frac{2S}{N_{c}}}{H\left( {{n/T_{s}},{{\mathbb{i}}\; T}} \right)}{u\left( {n,{\mathbb{i}}} \right)}{c_{PN}^{*}(n)}{w\left( {n,{\mathbb{i}}} \right)}} +}} \\{{\hat{n}\left( {n,{\mathbb{i}}} \right)}{c_{PN}^{*}(n)}{w\left( {n,{\mathbb{i}}} \right)}}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 15} \right\rbrack\end{matrix}$

That is, a multiplication result of the pseudo random-numbermultiplication portion 406 is further multiplied by w(n,i).

d{tilde over ( )}(n,i) obtained as above is so-called decision variable,and the detection portion 407 obtains an original signal (result ofencoding) from the decision variable according to the current modulationscheme.

Moreover, the parallel-serial conversion portion 409 conductsparallel-serial conversion and the decoder 410 conducts decoding so asto obtain a transmission signal.

Besides the above, the channel evaluation portion 408 examines whatinfluence the pilot symbol is subject to and sends feedback informationobtained by the influence to the transmission device 101 and also givesthe evaluation result by the channel evaluation portion 408 to thedetection portion 407.

In the above description, detailed description on the details ofadaptive modulation and methods of sending FBI, MLI is omitted, butvarious known arts can be used for that purpose.

However, as mentioned above, according to this embodiment, even if theFBI and MLI are sent by the unit of blocks, drop in performance issmall. This point is ascertained by experiments results, which will bedescribed later.

A method of determining the weight w(n,i) by the channel evaluationportion 408 will be further described below.

As shown in [Formula 13], in order to reduce the frequency distortioncaused by the frequency selective fading, frequency equalization andcombination is required. Here, a method of channel evaluation using Nppieces of pilot signals will be described.

The n-th channel response can be expressed as in [Formula 16]:

$\begin{matrix}{{\overset{\sim}{H}\left( {n/T_{s}} \right)} = {\frac{1}{N_{p}\sqrt{2{P/N_{c}}}}{\sum\limits_{i = 0}^{N_{p} - 1}{{\overset{\sim}{r}\left( {n,{\mathbb{i}}} \right)} \cdot {p^{*}\left( {n,{\mathbb{i}}} \right)} \cdot {c_{PN}^{*}({\mathbb{i}})}}}}} & \left\lbrack {{Formula}\mspace{14mu} 16} \right\rbrack\end{matrix}$

Here, for 0≦i≦Np,p(n,i)is a transmission pilot signal, and P is its intensity. The method ofdetermining the weight will be described below using this channelresponse H{tilde over ( )}(n/Ts).

(Method by ORC)

In ORC, the combined weight is made in inverse proportion to the channeltransfer function H(n/Ts) so as to fully restore the orthogonality.Therefore, the weight w_(ORC)(n,i) by the ORC is given by [Formula 17]:

$\begin{matrix}{{\omega_{ORC}\left( {n,{\mathbb{i}}} \right)} = \frac{1}{\overset{\sim}{H}\left( {n/T_{s}} \right)}} & \left\lbrack {{Formula}\mspace{14mu} 17} \right\rbrack\end{matrix}$

By using this weight, u(n,i) of the n-th sub carrier is obtained as in[Formula 18], [Formula 19]:

$\begin{matrix}\begin{matrix}{{\hat{u}\left( {n,{\mathbb{i}}} \right)} = {{\omega_{ORC}\left( {n,{\mathbb{i}}} \right)}{c_{PN}^{*}(n)}{\overset{\sim}{r}\left( {n,{\mathbb{i}}} \right)}}} \\{= {{\sqrt{\frac{2S}{N_{c}}}{\eta\left( {n,{\mathbb{i}}} \right)}{u\left( {n,{\mathbb{i}}} \right)}{c_{PN}^{*}(n)}} + \frac{{\hat{n}\left( {n,{\mathbb{i}}} \right)}{c_{PN}^{*}(n)}}{\overset{\sim}{H}\left( {n/T_{s}} \right)}}}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 18} \right\rbrack\end{matrix}$

$\begin{matrix}{{\eta\left( {n,{\mathbb{i}}} \right)} = \frac{H\left( {{n/T_{s}},{\mathbb{i}T}} \right)}{\overset{\sim}{H}\left( {n/T_{s}} \right)}} & \left\lbrack {{Formula}\mspace{14mu} 19} \right\rbrack\end{matrix}$

The decision variable d{tilde over ( )}(n,i) of the i-th data symbol ofthe n-th sub carrier is obtained as in [Formula 20]:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{d}\left( {n,{\mathbb{i}}} \right)} = {\sum\limits_{k = 0}^{N_{SF} - 1}{{\hat{u}\left( {{q + k},{\mathbb{i}}} \right)}{c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}}}} \\{= {\sum\limits_{k = 0}^{N_{SF} - 1}\left( {{\sqrt{\frac{2S}{N_{c}}}{\eta\left( {{q + k},{\mathbb{i}}} \right)}{u\left( {{q + k},{\mathbb{i}}} \right)}{c_{PN}^{*}\left( {q + k} \right)}} +} \right.}} \\{\left. \frac{\hat{n}\left( {{q + k},i} \right){c_{PN}^{*}\left( {q + k} \right)}}{\overset{\sim}{H}\left( {\left( {q + k} \right)/T_{s}} \right)} \right){c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}} \\{= {{\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{k = 0}^{N_{SF} - 1}{{\eta\left( {{q + k},{\mathbb{i}}} \right)}{d\left( {{q + k},{\mathbb{i}}} \right)}}}} +}} \\{\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{k = 0}^{N_{SF} - 1}{{\eta\left( {{q + k},{\mathbb{i}}} \right)}{{d_{intr}\left( {{q + k},{\mathbb{i}}} \right)} \cdot}}}} \\{{{c_{w}(k)}{c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}} +} \\{\sum\limits_{k = 0}^{N_{SF} - 1}\frac{{\hat{n}\left( {{q + k},{\mathbb{i}}} \right)}{c_{PN}^{*}(k)}{c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}}{\overset{\sim}{H}\left( {q + {k/T_{s}}} \right)}} \\{{{for}\mspace{14mu}\omega} \neq \left( {n\mspace{14mu}{mod}\mspace{14mu} N_{SF}} \right)}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 20} \right\rbrack\end{matrix}$

Here, q is floor(n/Nsf)·Nsf.

Referring to [Formula 20], it is known that the first term is a desiredsignal, the second term is an interference term, and the third term is anoise term.

From the third term, it is known that the orthogonality can be restoredby the ORC method, but it is also known that if the fading of the subcarrier is deep, the noise term becomes large.

(Method by MMSEC)

The combined weight w_(MMSEC)(n,i) in MMSEC is given by [Formula 21]:

$\begin{matrix}{{\omega_{MMSEC}\left( {n,{\mathbb{i}}} \right)} = \frac{\sqrt{\frac{2S}{N_{c}}} \cdot {\overset{\sim}{H}\left( {n/T_{s}} \right)}}{{{\sqrt{\frac{2S}{N_{c}}} \cdot {\overset{\sim}{H}\left( {n/T_{s}} \right)}}}^{2} + {2{\overset{\sim}{\sigma}}^{2}}}} & \left\lbrack {{Formula}\mspace{14mu} 21} \right\rbrack\end{matrix}$

Here, σ{tilde over ( )} is a noise intensity evaluated for each subcarrier, but in this embodiment, the noise intensity σ_(n){tilde over ()} in each sub carrier is supposed to be the same for all the subcarriers and to be σ{tilde over ( )}.

Here, the noise intensity σ_(n){tilde over ( )} of each sub carrier canbe acquired by [Formula 22]:

$\begin{matrix}{{\overset{\sim}{\sigma}}_{n}^{2} = {\frac{1}{N_{p}\sqrt{2{P/N_{c}}}}{\begin{matrix}{{\sum\limits_{i = 0}^{N_{p} - 1}{\overset{\sim}{r}\left( {n,{\mathbb{i}}} \right)}} -} \\{\sqrt{2{S/N}} \cdot {\overset{\sim}{H}\left( {n/T_{s}} \right)}}\end{matrix}}^{2}}} & \left\lbrack {{Formula}\mspace{14mu} 22} \right\rbrack\end{matrix}$

By supposition, it is σ_(n){tilde over ( )}²=σ{tilde over ( )}², and thenoise intensity σ{tilde over ( )} can be determined by [Formula 23]:

$\begin{matrix}{{\overset{\sim}{\sigma}}^{2} = {{\frac{1}{N_{c}}{\sum\limits_{n = 0}N_{c}}} - {1{\overset{\sim}{\sigma}}_{n}^{2}}}} & \left\lbrack {{Formula}\mspace{14mu} 23} \right\rbrack\end{matrix}$

At this time, the decision variable d{tilde over ( )}(n,i) of the i-thdata symbol of the n-th sub carrier can be expressed as in [Formula 24],[Formula 25]:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{d}\left( {n,{\mathbb{i}}} \right)} = {\sum\limits_{k = 0}^{N_{SF} - 1}{{\hat{u}\left( {{q + k},{\mathbb{i}}} \right)}{c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}}}} \\{= {\sum\limits_{k = 0}^{N_{SF} - 1}\left( {{\sqrt{\frac{2S}{N_{c}}}{\eta\left( {{q + k},{\mathbb{i}}} \right)}{u\left( {{q + k},{\mathbb{i}}} \right)}{c_{PN}^{*}\left( {q + k} \right)}} +} \right.}} \\{\left. \frac{\sqrt{\frac{2S}{N_{c}}}{\hat{n}\left( {{q + k},{\mathbb{i}}} \right)}{c_{PN}^{*}\left( {q + k} \right)}{\overset{\sim}{H}\left( {q + {k/T_{s}}} \right)}}{{{\sqrt{\frac{2S}{N_{c}}}{\overset{\sim}{H}\left( {q + {k/T_{s}}} \right)}}}^{2} + {2{\overset{\sim}{\sigma}}^{2}}} \right)c_{n\mspace{14mu}{mod}\mspace{14mu}{N_{SF}{(k)}}}^{*}} \\{= {{\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{k = 0}^{N_{SF} - 1}{{\lambda\left( {{q + k},{\mathbb{i}}} \right)}{d\left( {{q + k},{\mathbb{i}}} \right)}}}} +}} \\{{\sqrt{\frac{2S}{N_{c}}}{\sum\limits_{k = 0}^{N_{SF} - 1}{{\lambda\left( {{q + k},{\mathbb{i}}} \right)}{d_{intr}\left( {{q + k},{\mathbb{i}}} \right)}{c_{w}(k)}{c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}}}} +} \\{\sum\limits_{k = 0}^{N_{SF} - 1}\frac{\sqrt{\frac{2S}{N_{c}}}{\hat{n}\left( {{q + k},{\mathbb{i}}} \right)}{\overset{\sim}{H}\left( {q + {k/T_{s}}} \right)}{c_{PN}^{*}(k)}{c_{n\mspace{14mu}{mod}\mspace{14mu} N_{SF}}^{*}(k)}}{{{\sqrt{\frac{2S}{N_{c}}}{\overset{\sim}{H}\left( {q + {k/T_{s}}} \right)}}}^{2} + {2{\overset{\sim}{\sigma}}^{2}}}} \\{{{for}\mspace{14mu} w} \neq \left( {n\mspace{14mu}{mod}\mspace{14mu} N_{SF}} \right)}\end{matrix} & \left\lbrack {{Formula}\mspace{14mu} 24} \right\rbrack\end{matrix}$

$\begin{matrix}{{\lambda\left( {n,{\mathbb{i}}} \right)} = \frac{{H\left( {{n/T_{s}},{{\mathbb{i}}\; T}} \right)} \cdot {\overset{\sim}{H}\left( {n/T_{s}} \right)}}{{{\sqrt{\frac{2S}{N_{c}}}{\overset{\sim}{H}\left( {n/T_{s}} \right)}}}^{2} + {2{\overset{\sim}{\sigma}}^{2}}}} & \left\lbrack {{Formula}\mspace{14mu} 25} \right\rbrack\end{matrix}$

Here, it is q=floor(n/Nsf).Nsf.

(FBI and MLI)

FIG. 5 are explanatory diagrams illustrating a state of intensity of asub carrier. In FIG. 5A, a state in the case of AMS/OFDM by aconventional method is illustrated, while in FIG. 5B, a state in thecase of AMS/OFDM by the method of this embodiment is shown. Descriptionwill be made below referring to the figures.

Referring to these figures and [Formula 20], [Formula 23], it is knownthat in the same frequency equalization block, a desired signal,interference and noise/power ratio (SINR) are the same.

In the adaptive OFDM based on the frequency symbol diffusion, eachparallel signal is diffused on Nsf pieces of sub carriers by theorthogonal diffusion code with the length Nsf and then, combined.

Therefore, in each sub carrier, parallel signals with the same powerrate are superimposed.

In this case, even the sub carrier influenced by the frequency selectivefading as well as each parallel signal can obtain the same power rate.Therefore, the SINR of the detection signal becomes the same.

As a result, according to this embodiment, the same modulation level canbe assigned to each frequency symbol diffusion block 105.

Moreover, since the SINR of each sub carrier presents the same value inthe same frequency symbol diffusion block 105, the number of FBI and MLIrequired for each frequency symbol diffusion block 105 is 1, which isdifferent from the conventional AMS/OFDM.

Thus, according to this embodiment, the transmission amount of FBI andMLI can be reduced, and performance can be improved.

(Experiment Results)

The experiment results by a numerical simulation will be describedbelow. First, the following specification is used:

Modulation scheme is QPSK, 16QAM.

Demodulation is coherent detection.

Effective data rate is 20M symbols per second.

FFT size is 64.

The number of carriers is 64.

The guard interval length is 16 sample timing.

Frame size is 22 symbols (Np=2, Nd=20).

FEC is convolution code (rate R=½, restricted length K=7).

Fading is 7-path Rayleigh fading.

Doppler frequency is 10 Hz.

First, on the transmission side, data stream is encoded, and the aboveconvolution code is applied. This is known to be efficient fortransmitting an OFDM signal on the frequency selective fading channel.

Moreover, using AMC calculated by [Formula 20], [Formula 24], the codedbit is mapped to a modulation symbol of Nc pieces of sub carriers.

The modulation signal is serial-parallel converted, and each parallelsignal is diffused by the orthogonal diffusion code with the length Nsf(Walsh-Hadamart code and the like).

By this arrangement, each sub carrier has a plurality of parallel signalsuperimposed, and their power rates become the same.

An OFDM time signal is generated by inverse Fourier transform, and aftercyclic extension is inserted, it is transmitted on a frequencyselective/time change radio channel.

FIG. 6 is an explanatory diagram illustrating a state of transmissionchannel propagation to which the transmission signal is subject.Description will be made below referring to the figure.

A model shown in this figure has a shape in which path Rayleigh fadingwith L=7 is exponentially attenuated and has a path intervalT_(path=140ns).

In this case, the frequency selective fading can be a serious problem.

Suppose that the largest Doppler frequency is 10 Hz.

On the receiving side, the receiving signal is serial-parallelconverted, the parallel signal is fast-Fourier-transformed, and thesignal is returned to the frequency domain.

Since each signal is diffused by the orthogonal diffusion code on thetransmission side, each signal can be detected by the orthogonaldiffusion code.

However, the orthogonality between different diffusion codes might belost by the frequency selective fading.

Then, using the frequency equalization and combination technology, theorthogonality is restored. In this simulation, the ORC method and theMMSEC method are employed as the equalization method.

The signal equalized as above is demodulated by the decision variableobtained according to [Formula 20], [Formula 24].

After the demodulation, binary data is decoded by Viterbi soft-decodingalgorithm.

FIG. 7 is an explanatory diagram illustrating a packet structure.Description will be made below referring to this figure.

The packet comprises 64 sub carriers and 22 OFDM symbols. The number ofpilot symbols Np is 2, and the number of data Nd is 20. Duration of asingle OFDM symbol is 11.2 μs.

FIG. 8 is a graph illustrating the BER value to the conventional OFDMand the BER value to the FSS-OFDM using ORC and MMSEC. Description willbe made below referring to the figure.

The BER of the FFS-OFDM using ORC is poorer than the conventional OFDMat low E_(b)/N₀. That is because a noise is generated even in a statewithout an error in the ORC-based FFS-OFDM system.

On the other hand, the MMSEC method generates the best BER performanceand that is because power loss is minimized while influence of noise isrestricted using all the sub carriers.

FIG. 9 is a graph showing the BER values of FFS-OFDM when Nsf=2, 4, 16,64 in MMSEC. Description will be made below referring to this figure.

As shown in this figure, the larger Nsf becomes, the better the BER isimproved. That is because frequency diversity is carried out inFFS-OFDM. If Nsf is small, correlation of the subsequent sub carrierbecomes stronger, and a degree of the frequency diversity is lowered. Inthis way, the degree of the frequency diversity can be increased bylarge Nsf in FFS-OFDM.

However, if Nsf is made too large, a diffusion band width would be widerthan the coherent band width.

FIG. 10 shows the BER of FFS-OFDM using the convolution code for variousNsf for ORC and MMSEC. Description will be made below referring to thisfigure.

As shown in this figure, various BER is obtained at different Nsf.

However, by using FEC and interleave, if Nsf is the same, the BER isconsidered to present approximately the same performance. Therefore, thefrequency diversity can be sufficiently conducted by using FEC andinterleave.

FIG. 11 is a graph illustrating throughputs of fixed QPSK OFDM, fixed16QAM OFDM, conventional AMS/OFDM, AMS/FSC-OFDM with ORC, andAMS/FSC-OFDM with MMSEC. Description will be made below referring to thefigure.

In the AMS/FSC-OFDM with ORC, AMS/FSC-OFDM with MMSEC according to thisembodiment, only one SINR is required as FBI for appropriate modulation,and this is different from fixed QPSK OFDM, fixed 16QAM OFDM,conventional AMS/OFDM according to the conventional method.

Therefore, the system of this embodiment has the best throughputperformance.

On the other hand, in the conventional AMS/OFDM system, MLI istransmitted as data, and the transmission rate is lower than that of thesystem of this embodiment.

FIG. 12 is a graph illustrating the transmission amounts of FBI and MLIof the conventional AMS/OFDM and AMS/FSS-OFDM of this embodiment forNsf=4, 16, 64. Description will be made below referring to the figure.

As shown in this figure, when the transmission amounts of FBI and MLI atNsf=64 in AMS/FSS-OFDM of this embodiment is α, the transmission amountat Nsf=16 in AMS/FSS-OFDM of this embodiment is 4α, the transmissionamount at Nsf=4 in AMS/FSS-OFDM of this embodiment is 16α, and thetransmission amount of the conventional AMS/OFDM is approximately 64α

Therefore, the transmission amount α of FBI and MLI of AMS/FSS-OFDM ofthis embodiment is considerably small.

INDUSTRIAL APPLICABILITY

As mentioned above, according to the present invention, the transmissiondevice, receiving device, transmission method, receiving method,computer-readable information recording medium recording a program forrealizing them using a computer, and the program, which are suitable forrealizing improvement of performance of adaptive OFDM, can be provided.

1. A transmission device comprising: a serial-parallel conversionportion that serial-parallel converts a transmission signal to aplurality of signals corresponding to Nc number of subcarriers andoutputs the plurality of signals, the i-th symbol in the time directionof the n-th signal in the plurality of signals being:d(n,i); a frequency symbol diffusion portion that outputs a plurality ofsignals with substantially the same transmitting electrical power withrespect to the output plurality of signals d(n,i) for each signal group,in which each Nsf signals in order of n of the plurality of signalsd(n,i) is grouped, using a plurality of complex orthogonal diffusionseriesc₀(0),c₀(1), . . . ,c₀(Nsf−1),c₁(0),c₁(1), . . . ,c₁(Nsf−1),. . .c_(Nsf−1)(0),c_(Nsf−1)(0), . . . ,c_(Nsf−1)(Nsf−1), with a length of Nsfwith respect to said outputted plurality of signals d(n,i), wherein|c _(k)(m)|=1and if k=w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=Nsf;if k≠w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=0 and an expression (·)* acquirescomplex conjugation and floor (·) conducts truncation, among theplurality of signals, the i-th symbol in the time direction of the n-thsignal is:u(n,i)=Σ_(k=0) ^(Nsf−1) c _(k)(n mod Nsf)·d(floor(n/Nsf)·Nsf+k,i); apseudo random-number multiplication portion that multiplies each of saidoutputted plurality of signals u(n,i) by a pseudo random-number codeseriesc_(PN)(n) out of the pseudo random-number code seriesc_(PN)(0),c_(PN)(1), . . . and outputs the result; an inverse Fouriertransform portion that conducts inverse Fourier transform of saidoutputted plurality of signalsc_(PN)(n)·u(n,i) and outputs a plurality of signals; a parallel-serialconversion portion that parallel-serial converts said plurality ofsignals outputted after the inverse Fourier transform; and atransmission portion that transmits the signal of the result of saidparallel-serial converted signal; and an adaptive modulating portionthat adaptive modulates the transmission signal on the basis of feedbackinformation for each signal group transmitted from a receiving device.2. A receiving device communicating with the transmission deviceaccording to claim 1, the receiving device comprising: a receivingportion that receives a signal transmitted from said transmissiondevice; a serial-parallel conversion portion that serial-parallelconverts said received signal to Nc number of groups and outputs aplurality of signals; a Fourier transform portion that conducts Fouriertransform of said plurality of serial-parallel converted and outputtedsignals and outputs a plurality of signals, wherein the i-th symbol inthe time direction of the n-th signal among said plurality ofFourier-transformed and outputted signals is:r{tilde over ( )}(n,i); a pseudo random-number multiplication portionthat multiplies each of said plurality of Fourier-transformed andoutputted signals r{tilde over ( )}(n,i) by complex conjugationc_(PN)(n)*ofc_(PN)(n) among the pseudo random-number code series and outputs it; aweight calculation portion that calculates a weight to the i-th symbolof the n-th signal:w(n,i); a detection portion that multiplies the plurality of signalsmultiplied by the complex conjugation c_(PN)(n)* and outputted by saidcalculated weight w(n,i) and outputs a plurality of signals:u^(n,i)=w(n,i)·c _(PN)(n)*·r{tilde over ( )}(n,i); a frequencyequalization and combination portion that performs frequencyequalization and combination to said outputted plurality of signalsu(n,i) and outputs a plurality of signals, wherein the i-th symbol inthe time direction of the n-th signal in the plurality of signals is:d{tilde over ( )}(n,i)=Σ_(k=0) ^(Nsf−1) u(floor(n/Nsf)·Nsf+k,i)·c_(n mod Nsf)(k)*; and a parallel-serial conversion portion thatparallel-serial converts said outputted plural signals d{tilde over ()}(n,i) and obtains a transmission signal: a channel evaluating portionfor creating feedback information for each signal group, in which eachset of Nsf signals in the order of n of the plurality of signals d{tildeover ( )}(n,i) outputted is grouped, and for transmitting the feedbackinformation to the transmitting device.
 3. The receiving deviceaccording to claim 2, further comprising a channel transfer functioncalculation portion that calculates, using a pilot signal p(n,i) withintensity P, length Np and the effective symbol length Ts transmittedfrom said transmission device, a channel transfer function H{tilde over( )}(n/Ts) by:H{tilde over ( )}(n/Ts)=1/(Np·(2P/Nc)^(1/2))Σ_(i=0) ^(Np−1) r{tilde over( )}(n,i)·p(n,i)*·c _(PN)(i)*, wherein the weight w(n,i) is determinedfrom the channel transfer function H{tilde over ( )}(n/Ts).
 4. Thereceiving device according to claim 3, wherein the weight w(n,i) isdetermined as:w(n,i)=1/H{tilde over ( )}(n/Ts).
 5. The receiving device according toclaim 3, wherein by an average σ{tilde over ( )}² of noise intensityevaluated for each of the plurality of signals r{tilde over ( )}(n,i),the weight w(n,i) is determined, with respective to an averagetransmitting electrical power S, as:w(n,i)=(2S/Nc)^(1/2) ·H{tilde over ( )}(n/Ts)/(|(2S/Nc)^(1/2) ·H{tildeover ( )}(n/Ts)|²+2σ²).
 6. A transmission method comprising steps of:serial-parallel conversion of serial-parallel converting a transmissionsignal to a plurality to signals corresponding to Nc number ofsubcarriers and outputting the plurality of signals, wherein the i-thsymbol in the time direction of the n-th signal in the plurality ofsignals is:d(n,i); a frequency symbol diffusion that outputs a plurality of signalswith substantially the same transmitting electrical power with respectto the output plurality of signals d(n,i) for each signal group, inwhich each Nsf signals in order of n of the plurality of signals d(n,i)is grouped, using a plurality of complex orthogonal diffusion seriesc₀(0),c₀(1), . . . ,c₀(Nsf−1),c₁(0),c₁(1), . . . ,c₁(Nsf−1),. . .c_(Nsf−1)(0),c_(Nsf−1)(1), . . . ,c_(Nsf−1)(Nsf−1), with the length ofNsf with respect to said outputted plurality of signals d(n,i), wherein|c _(k)(m)|=1and if k=w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=Nsf;if k≠w,Σ_(m=0) ^(Nsf−1) c _(k)(m)·c _(w)(m)*=0 and an expression (·)* acquirescomplex conjugation and floor (·) conducts truncation, among theplurality of signals, the i-th symbol in the time direction of the n-thsignal is:u^(n,i)Σ_(k=0) ^(Nsf−1) c _(k)(n mod Nsf)·d(floor(n/Nsf)·Nsf+k,i);pseudo random-number multiplication of multiplying each of saidoutputted plurality of signals u(n,i) by the pseudo random-number codeseriesc_(PN)(n) out of the pseudo random-number code seriesc_(PN)(0),c_(PN)(1), . . . and outputting the result; conducting inverseFourier transform of said outputted plurality of signalsc_(PN)(n)·u(n,i) and outputting a plurality of signals; parallel-serialconverting said inverse-Fourier-transformed and outputted plurality ofsignals; transmitting the signal of the result of said parallel-serialconversion; and adaptive modulating for adaptive modulating thetransmission signal on the basis of feedback information for each signalgroup.
 7. A receiving method for receiving a signal by the transmissionmethod according to claim 6, comprising steps of: receiving a signaltransmitted by said transmission method; serial-parallel converting saidreceived signal to Nc number of groups and outputting a plurality ofsignals; conducting Fourier transform of said plurality ofserial-parallel converted and outputted signals and outputting aplurality of signals, wherein the i-th symbol in the time direction ofthe n-th signal among said plurality of Fourier-transformed andoutputted signals is:r{tilde over ( )}(n,i) pseudo random-number multiplication ofmultiplying each of said plurality of Fourier-transformed and outputtedsignals r{tilde over ( )}(n,i) by complex conjugationc_(PN)(n)*ofc_(PN)(n)* among the pseudo random-number code series and outputting it;calculating a weight to the i-th symbol of the n-th signal:w(n,i); detection in which the plurality of signals multiplied by thecomplex conjugation c_(PN)(n)* and outputted by are multiplied by saidcalculated weight w(n,i) and a plurality of signals are outputted:u(n,i)=w(n,i)·c _(PN)(n)*·r{tilde over ( )}(n,i) frequency equalizationand combination in which frequency equalization and combination isperformed to said outputted plurality of signals u(n,i) and a pluralityof signals are outputted, wherein the i-th symbol in the time directionof the n-th signal in the plurality of signals is:d{tilde over ( )}(n,i)=Σ_(k=0) ^(Nsf−1) u(floor(n/Nsf)·Nsf+k,i)·c_(n mod Nsf)(k)*; and parallel-serial conversion of parallel-serialconverting said outputted plural signals d{tilde over ( )}(n,i) so as toobtain a transmission signal; and a channel evaluating for creatingfeedback information for each signal group, in which each set of Nsfsignals in the of n of the plurality of signals d{tilde over ( )}(n,i)outputted is grouped, and for transmitting the feedback information. 8.The receiving method according to claim 7, further comprising a step ofchannel transfer function calculation, using a pilot signal p(n,i) withintensity P, length Np and an effective symbol length Ts transmitted bysaid transmission method, the channel transfer function H{tilde over ()}(n/Ts) being calculated by:H{tilde over ( )}(n/Ts)=1/(Np·(2P/Nc)^(1/2))Σ_(i=0) ^(Np−1) r{tilde over( )}(n,i)·p(n,i)*·c _(PN)(i)*, wherein the weight w(n,i) is determinedfrom the channel transfer function H{tilde over ( )}(n/Ts).
 9. Thereceiving method according to claim 8, wherein the weight w(n,i) isdetermined as:w(n,i)=1/H{tilde over ( )}(n/Ts).
 10. The receiving method according toclaim 8, wherein by an average σ{tilde over ( )}² of noise intensityevaluated for each of the plurality of signals r{tilde over ( )}(n,i),the weight w(n,i) is determined, with respective to an averagetransmitting electrical power S, as:w(n,i)=(2S/Nc)^(1/2) ·H{tilde over ( )}(n/Ts)/(|(2S/Nc)^(1/2) ·H{tildeover ( )}(n/Ts)|²+2σ²).